Power converter with reduced root mean square input current

ABSTRACT

A power converter includes a network of switches having a first capacitor switch coupled to a first flying-capacitor, a first switch to couple a second flying-capacitor to a first port, an inductor coupled to a ground switch. A driver drives the network of switches in two states. In the first state the ground port is coupled to a second port via a first path comprising the first flying-capacitor and the inductor, and the first port is coupled to the second port via a second path comprising the first switch, the second flying-capacitor and the inductor. In the second state the ground is coupled to the second port via a third path comprising the ground switch and the inductor, and one of the first port and the ground port is coupled to the second port via a fourth path comprising the first flying-capacitor while bypassing the inductor.

RELATED PATENT APPLICATION

This application is related to DS20-001G, U.S. application Ser. No.16/900,669, now issued as U.S. Pat. No. 11,228,243, filed on Jun. 12,2020, the same day as the instant application, which is assigned to acommon assignee, and which is herein incorporated by reference in itsentirety.

TECHNICAL FIELD

The present disclosure relates to a power converter and a method ofoperating the same. In particular, the present disclosure relates to apower converter operable with a small output-to-input voltage conversionratio, for instance a conversion ratio Vout/Vin<1/4.

BACKGROUND

Typical Voltage-Regulator-Modules (VRMs) such as the ones used inindustrial, server, networking and computing applications are suppliedfrom a supply voltage (e.g. 12V) that is much higher than the maximuminput voltage of the load. For instance, the voltage supply may be 12Vand the input voltage of the load may be <1.8V for CPU, GPU, SoC orother memory module. There is therefore a need for efficient powerconversion when the output-to-input voltage ratio V_(OUT)/V_(IN) isrelatively small for instance less than 1/4.

Traditional buck converter and 3-level buck converter implement astep-down voltage conversion by pulling (for a duty cycle D<1) a currentpulse from the input terminal equal to the load current. As a result,the amplitude of input current pulses is equal to the load currentresulting in application noise and EMI issues. Multi-phase convertersdistribute the load current across multiple inductors, which increasesthe duty cycle during which a current is pulled from the input terminal.Therefore, the level of the switched input current is reduced by thenumber of phases up to certain duty cycle. However, this approachrequires the use of multiple coils that increase the total inductor coreloss.

Besides, in conventional DC-DC converters the inductor acts for shorttime intervals as a constant current source. Consequently, the inductorrequires a significant time to respond to a sudden change in loadcurrent, hence providing limited transient load response.

It is an object of the disclosure to address one or more of theabove-mentioned limitations.

SUMMARY

According to a first aspect of the disclosure, there is provided a powerconverter for providing an output voltage with a target conversionratio, the power converter having a ground port, a first port, and asecond port, wherein when the power converter operates as a step-downconverter the first port receives an input voltage and the second portprovides the output voltage and when the power converter operates as astep-up converter the second port receives an input voltage and thefirst port provides the output voltage; the power converter comprising afirst flying capacitor coupled to a network of switches, an inductorcoupled to the second port, and a driver; the network of switchescomprising a first switch to couple the first flying capacitor to thefirst port; a second switch to couple the inductor to ground; the driverbeing adapted to drive the network of switches with a sequence of statesduring a drive period, the sequence of states comprising a first stateand a second state, wherein in the first state the ground port iscoupled to the second port via a first path comprising the first flyingcapacitor and the inductor, and the first port is decoupled from thesecond port, wherein in the second state the ground port is coupled tosecond port via a second path comprising the second switch and theinductor, and wherein the first port is coupled to the second port via athird path comprising the first flying capacitor while bypassing theinductor.

Optionally, the power converter further comprises a second flyingcapacitor coupled to the second port via a first inductor switch,wherein the network of switches comprises a capacitor switch between thefirst flying capacitor and the second flying capacitor.

Optionally, wherein in the first state the first path comprises thefirst flying capacitor, the capacitor switch, the second flyingcapacitor and the inductor, and wherein in the second state the groundterminal is coupled to the second port via a ground path comprising thesecond switch, the second flying capacitor and the first inductorswitch, while bypassing the inductor.

Optionally, the network of switches comprises a second capacitor switchbetween the first flying capacitor and the second flying capacitor, anda ground switch to couple the second flying capacitor to ground.

Optionally, the inductor is coupled to the first flying capacitor via asecond inductor switch, and the first flying capacitor is coupled to thesecond port via a third capacitor switch.

Optionally, wherein in the first state the ground port is coupled to thesecond port via another path comprising the ground switch, the secondflying capacitor, the first inductor switch and the inductor; andwherein in the second state the first port is coupled to the second portvia a path comprising the capacitor switch, the second flying capacitor,the second capacitor switch, the first flying capacitor and the thirdcapacitor switch, while bypassing the inductor.

Optionally, the power converter further comprises a third flyingcapacitor, the third flying capacitor having a first terminal coupled tothe first flying capacitor via a first coupling switch, and a secondterminal coupled to the first flying capacitor via a second couplingswitch.

Optionally, the driving sequence comprises a primary first-state, asecondary first-state, a primary second-state and a secondarysecond-state.

Optionally, wherein in the primary second-state the ground port iscoupled to the second port via a first ground path comprising the secondflying capacitor while bypassing the inductor, and wherein the firstport is coupled to the second port via a path comprising the first andthird flying capacitors while bypassing the inductor.

Optionally, wherein in the secondary second state the ground port iscoupled to the second port via the first ground path, and a secondground path comprising the first and third flying capacitors whilebypassing the inductor.

Optionally, wherein in the primary and secondary first states the groundport is coupled to the second port via a path comprising the firstflying capacitor, the second flying capacitor and the inductor.

Optionally, the driving sequence comprises a first additional state, anda second additional state, wherein in the first additional state thefirst port is coupled to the second port via a path comprising thesecond and third flying capacitors and the inductor, and wherein in thesecond additional state the ground port is coupled to the second portvia a path comprising the second and third flying capacitors and theinductor.

Optionally, the driving sequence comprises another first state in whichthe ground port is de-coupled from the second port and wherein the firstport is coupled to the second port via a path comprising the inductor.

Optionally, the power converter further comprises a current sensor forsensing an inductor current through the inductor, wherein the driver isadapted to open the second switch during the second state upon sensingthat the inductor current has reached a threshold value. For instance,the threshold value may be a null current value.

Optionally, the power converter is a step-down converter, the firststate being a magnetization state and the second state being ade-magnetization state.

Optionally, the power converter is a step-up converter, the first statebeing a de-magnetization state and the second state being amagnetization state.

According to a second aspect of the disclosure, there is provided amethod of converting power with a target conversion ratio, the methodcomprising providing a power converter having a ground port, a firstport, and a second port, wherein when the power converter operates as astep-down converter the first port receives an input voltage and thesecond port provides the output voltage and when the power converteroperates as a step-up converter the second port receives an inputvoltage and the first port provides the output voltage; the powerconverter comprising a first flying capacitor coupled to a network ofswitches, an inductor coupled to the second port, and a driver; thenetwork of switches comprising a first switch to couple the first flyingcapacitor to the first port; a second switch to couple the inductor toground; driving the network of switches with a sequence of states duringa drive period, the sequence of states comprising a first state and asecond state, wherein in the first state the ground port is coupled tothe second port via a first path comprising the first flying capacitorand the inductor, and the first port is decoupled from the second port,wherein in the second state the ground port is coupled to second portvia a second path comprising the second switch and the inductor, andwherein the first port is coupled to the second port via a third pathcomprising the first flying capacitor while bypassing the inductor.

The options described with respect to the first aspect of the disclosureare also common to the second aspect of the disclosure.

According to a third aspect of the disclosure there is provided a powerconverter for providing an output voltage with a target conversionratio, the power converter having a ground port, a first port, and asecond port, wherein when the power converter operates as a step-downconverter the first port receives an input voltage and the second portprovides the output voltage and when the power converter operates as astep-up converter the second port receives an input voltage and thefirst port provides the output voltage, the power converter comprising afirst flying capacitor coupled to a network of switches, a second flyingcapacitor coupled to the network of switches, an inductor coupled to thesecond port, and a driver; the network of switches comprising a firstswitch to couple the second flying capacitor to the first port; a groundswitch to couple the inductor to ground; a first capacitor switchcoupled to the first flying capacitor; the driver being adapted to drivethe network of switches with a sequence of states during a drive period,the sequence of states comprising a first state and a second state,wherein in the first state the ground port is coupled to the second portvia a first path comprising the first flying capacitor and the inductor,and wherein the first port is coupled to the second port via a secondpath comprising the first switch, the second flying capacitor and theinductor, wherein in the second state the ground port is coupled to thesecond port via a third path comprising the ground switch and theinductor, and wherein one of the first port and the ground port iscoupled to the second port via a fourth path comprising the first flyingcapacitor while bypassing the inductor.

Optionally, in the second state the first port is decoupled from thesecond port and wherein the fourth path comprises the second flyingcapacitor.

Optionally, in the second state the ground port is coupled to the secondport via a fifth path comprising the second flying capacitor, whilebypassing the inductor.

Optionally, the inductor has a first terminal coupled to the firstflying capacitor via a first inductor switch, and a second terminalconnected to the second port, and wherein the first flying capacitor iscoupled to the second port via a second capacitor switch.

Optionally, the second flying capacitor is coupled to the second portvia a third capacitor switch.

Optionally, the power converter further comprises a current sensor forsensing an inductor current through the inductor, wherein the driver isadapted to open the ground switch during the second state upon sensingthat the inductor current has reached a threshold value. For instance,the threshold value may be a zero current value.

Optionally, the power converter is a step-down converter, the firststate being a magnetization state and the second state being ade-magnetization state.

Optionally, the power converter is a step-up converter, the first statebeing a de-magnetization state and the second state being amagnetization state.

According to a fourth aspect of the disclosure, there is provided amethod of converting power with a target conversion ratio, the methodcomprising providing a power converter having a ground port, a firstport, and a second port, wherein when the power converter operates as astep-down converter the first port receives an input voltage and thesecond port provides the output voltage and when the power converteroperates as a step-up converter the second port receives an inputvoltage and the first port provides the output voltage, the powerconverter further comprising a first flying capacitor coupled to anetwork of switches, a second flying capacitor coupled to the network ofswitches, an inductor coupled to the second port, and a driver; whereinthe network of switches comprises a first switch to couple the secondflying capacitor to the first port; a ground switch to couple theinductor to ground; a first capacitor switch coupled to the first flyingcapacitor; driving the network of switches with a sequence of statesduring a drive period, the sequence of states comprising a first stateand a second state, wherein in the first state the ground port iscoupled to the second port via a first path comprising the first flyingcapacitor and the inductor, and wherein the first port is coupled to thesecond port via a second path comprising the first switch, the secondflying capacitor and the inductor, wherein in the second state theground port is coupled to the second port via a third path comprisingthe ground switch and the inductor, and wherein one of the first portand the ground port is coupled to the second port via a fourth pathcomprising the first flying capacitor while bypassing the inductor.

Optionally, in the second state the first port is decoupled from thesecond port and wherein the fourth path comprises the second flyingcapacitor.

Optionally, in the second state the ground port is coupled to the secondport via a fifth path comprising the second flying capacitor, whilebypassing the inductor.

The options described with respect to the third aspect of the disclosureare also common to the fourth aspect of the disclosure.

BRIEF DESCRIPTION OF THE DRAWINGS

The disclosure is described in further detail below by way of exampleand with reference to the accompanying drawings, in which:

FIG. 1A is a diagram of a two-level Buck converter;

FIG. 1B is a diagram of a three-level Buck converter;

FIG. 1C is a diagram of a combined Buck converter and capacitivedivider;

FIG. 1D is a diagram of a multiphase Buck converter;

FIG. 1E is a diagram of another multiphase Buck converter;

FIG. 2 is a flow chart of a method for providing a voltage with a inputto output conversion ratio according to the disclosure;

FIG. 3 is a diagram of a power converter for implementing the method ofFIG. 2;

FIG. 4A is a diagram of a magnetization state for operating the powerconverter of FIG. 3;

FIG. 4B is a diagram of a de-magnetization state for operating the powerconverter of FIG. 3;

FIG. 4C is a diagram of another magnetization state for operating thepower converter of FIG. 3;

FIG. 4D is a diagram of another de-magnetization state for operating thepower converter of FIG. 3;

FIG. 5 is a plot of a drive sequence for operating the power converterof FIG. 3 with a specific conversion ratio;

FIG. 6 is a diagram of another power converter for implementing themethod of FIG. 2;

FIG. 7A is a diagram of a magnetization state for operating the powerconverter of FIG. 6;

FIG. 7B is a diagram of a de-magnetization state for operating the powerconverter of FIG. 6;

FIG. 7C is a diagram of another magnetization state for operating thepower converter of FIG. 6;

FIG. 8 is a diagram of another power converter for implementing themethod of FIG. 2;

FIG. 9A is a diagram of a magnetization state for operating the powerconverter of FIG. 8;

FIG. 9B is a diagram of a de-magnetization state for operating the powerconverter of FIG. 8;

FIG. 10 is a diagram of another power converter for implementing themethod of FIG. 2;

FIG. 11A is a diagram of a first magnetization state for operating thepower converter of FIG. 10;

FIG. 11B is a diagram of a first de-magnetization state for operatingthe power converter of FIG. 10;

FIG. 11C is a diagram of a second magnetization state for operating thepower converter of FIG. 10;

FIG. 11D is a diagram of a second de-magnetization state for operatingthe power converter of FIG. 10;

FIG. 12 is a plot of a drive sequence for operating the power converterof FIG. 10;

FIG. 13A is a diagram of another magnetization state for operating thepower converter of FIG. 10;

FIG. 13B is a diagram of yet another magnetization state for operatingthe power converter of FIG. 10.

FIG. 14 is a flow chart of another method for providing a voltage withan input to output conversion ratio according to the disclosure;

FIG. 15 is a diagram of a power converter for implementing the method ofFIG. 14;

FIG. 16A is a diagram of a magnetization state for operating the powerconverter of FIG. 15;

FIG. 16B is a diagram of a de-magnetization state for operating thepower converter of FIG. 15;

FIG. 17 is a diagram of another power converter for implementing themethod of FIG. 14;

FIG. 18A is a diagram of a magnetization state for operating the powerconverter of FIG. 17;

FIG. 18B is a diagram of a de-magnetization state for operating thepower converter of FIG. 17

FIG. 19 is a diagram of FIG. 3 represented with inverted input andoutput ports;

FIG. 20A is a diagram of a magnetization state for operating the powerconverter of FIG. 19;

FIG. 20B is a diagram of a de-magnetization state for operating thepower converter of FIG. 19;

FIG. 21 is a diagram of another de-magnetization state for operating thepower converter of FIG. 19.

DETAILED DESCRIPTION

FIGS. 1A and 1B show the topologies of traditional two-levels andthree-levels Buck converters. The two-level Buck converter provides anoutput current alternatively from the input terminal and the groundterminal. Consequently, the level of pulsed input current I_(IN) (duringinductor magnetization) is equal to the load current I_(OUT) (zerootherwise):

$\begin{matrix}\begin{matrix}{\frac{I_{IN}}{I_{OUT}} = 1} & {D \in \left\lbrack {0,1} \right\rbrack}\end{matrix} & (1)\end{matrix}$

For the 3-level Buck converter, the flying capacitor C_(F) may beregulated to V_(CF)=V_(IN)/2 so that the magnetization voltage acrossthe inductor L is reduced towards V_(L)=V_(IN)/2−V_(OUT).

FIG. 1C illustrates a combined Buck converter and capacitive voltagedivider according to U.S. Pat. No. 8,427,113. In this example the flyingcapacitor C_(F) and the reservoir capacitor C_(R) are automaticallycharged to the same voltage V_(CF)=V_(CR)=V_(IN)/2. During the inductormagnetization state, the load current is supplied in parallel by theinput terminal and the (charged) reservoir capacitor C_(R), hencereducing the amplitude of input current pulses (discontinuous current)to 1/2 of the load current:

$\begin{matrix}\begin{matrix}{\frac{I_{IN}}{I_{OUT}} = \frac{1}{2}} & {D \in \left\lbrack {0,1} \right\rbrack}\end{matrix} & (2)\end{matrix}$

The above result describes a ratio of average currents during the periodof inductor magnetization (neglecting impact from inductor currentripple). The relationship between input and output voltages is obtainedby applying the volt-sec balance principle to the voltage of theinductor during the inductor magnetization switching state DP and theinductor demagnetization switching state DV:

$\begin{matrix}\begin{matrix}{\frac{V_{OUT}}{V_{IN}} = \frac{D}{2}} & {{{DP} = D},{{DV} = {1 - D}}} & {D \in \left\lbrack {0,1} \right\rbrack}\end{matrix} & (3)\end{matrix}$

From equation (3) one can derive a theoretical maximum voltageconversion ratio a V_(OUT)/V_(IN)=1/2 for D=1, in which D is the dutycycle of the inductor magnetization state that connects the input to theoutput port of the converter. However, for D=1 there is zero timeavailable to re-distribute the charge from flying capacitor C_(F) intothe reservoir capacitor C_(R) as this would require an infinite currentcausing a corresponding infinite I²R conduction loss. A more balancedcurrent distribution is achieved by restricting the duty cycle to avalue smaller than 1, for instance D≤3/4, resulting into a practicalmaximum voltage conversion ratio of V_(OUT)/V_(IN)=3/8 for D=3/4.

FIG. 1D shows a diagram of a hybrid multiphase Buck converter, alsoreferred to as a series capacitor Buck converter according to U.S. Pat.No. 7,230,405. In this example, during the magnetization (from the inputterminal) of one inductor, half of the load current is provided via thesecond inductor (demagnetized from the ground terminal). As a result theamplitude of the current pulses generated at the input is reduced.

$\begin{matrix}\begin{matrix}{\frac{I_{IN}}{I_{OUT}} = \frac{1}{2}} & {D \in \left\lbrack {0,0.5} \right\rbrack}\end{matrix} & (4)\end{matrix}$

The flying capacitor may be regulated to V_(CF)=V_(IN)/2, so that therelation between input and output voltages follows equation (3).However, for a balanced inductor load current the maximum possible dutycycle is reduced to D=0.5:

$\begin{matrix}\begin{matrix}{\frac{V_{OUT}}{V_{IN}} = \frac{D}{2}} & {{{DP} = D},{{DV} = {1 - D}}} & {D \in \left\lbrack {0,{0.5}} \right\rbrack}\end{matrix} & (5)\end{matrix}$

This corresponds to a maximum voltage conversion ratio ofV_(OUT)/V_(IN)=1/4 for D=0.5.

FIG. 1E illustrates a derivative topology of the converter of FIG. 1D inwhich the two flying capacitors C1 and C2 are both regulated toV_(C1)=V_(C2)=V_(IN)/2, hence reducing the amplitude of input currentpulses further down to:

$\begin{matrix}\begin{matrix}{\frac{I_{IN}}{I_{OUT}} = \frac{1}{4}} & {D \in \left\lbrack {0,0.5} \right\rbrack}\end{matrix} & (6)\end{matrix}$

FIG. 2 is a flow chart of a method for converting power with a targetconversion ratio according to the disclosure.

At step 210 a power converter having a ground port, a first port, and asecond port is provided. The power converter can operate either as astep-down converter or as a step-up converter. When the power converteroperates as a step-down converter the first port receives an inputvoltage and the second port provides the output voltage. When the powerconverter operates as a step-up converter the second port receives aninput voltage and the first port provides the output voltage. The powerconverter includes a first flying capacitor coupled to a network ofswitches, an inductor coupled to the second port, and a driver. Thenetwork of switches comprises a first switch to couple the first flyingcapacitor to the first port; a second switch to couple the inductor toground.

At step 220 the network of switches is driven with a sequence of statesthat include a first state and a second state. In the first state theground port is coupled to the second port via a first path comprisingthe first flying capacitor and the inductor, and the first port isdecoupled from the second port. In the second state the ground port iscoupled to second port via a second path comprising the second switchand the inductor, and wherein the first port is coupled to the secondport via a third path comprising the first flying capacitor whilebypassing the inductor. As a result, in the first state there is nocurrent flowing between the first port and the second port.

When the power converter operates as a step-down converter, the firststate is a magnetization state and the second state is ade-magnetization state. Conversely, when the power converter operates asa step-up converter, the first state is a de-magnetization state and thesecond state is a magnetization state.

Optionally, a current sensor may be provided for sensing an inductorcurrent through the inductor. Then the second switch may be openedduring the second state upon sensing that the inductor current hasreached a threshold value. This permits to disable current flowing fromthe output towards ground (negative inductor current) at low outputcurrent.

Using the method of FIG. 2 permits to deliver efficient power conversionespecially for small step-down output-to-input voltage conversion ratio,for instance for V_(OUT)/V_(IN)<1/4, or for large step-up voltageconversion ratios. By implementing a capacitive current path bypassingthe inductor, the losses due to the inductor DC resistance can bereduced hence improving converter efficiency and voltage regulation. Inaddition, the flying capacitor may act as supplement output capacitance,hence improving the response to transient load current.

FIG. 3 is a diagram of a DC-DC converter 300 for implementing the methodof FIG. 2. The DC-DC converter 300 includes an inductor L and a flyingcapacitor C_(F) coupled between a first port (input node 302) and asecond port (output node 304) by a network of switches formed of fiveswitches S1, S2, S3, S4, S5. An input capacitor Cin is provided betweenthe input node 302 and ground and an output capacitor Cout is providedbetween the output node 304 and ground. The capacitors Cin and Cout areconnected to a fixed ground voltage and may be referred to as reservoircapacitors. The capacitor C_(F) has terminals provided with varyingvoltages and may be referred to as a flying capacitor.

The flying capacitor C_(F) is coupled to the input node 302 via a firstswitch, the input switch S1, and to ground via the ground switch S4. Theflying capacitor C_(F) has a first terminal coupled to node 306 and asecond terminal coupled to node 308. In addition, the second terminal ofcapacitor C_(F) is coupled to the output node 304 via the switch S3. Theinductor L has a first terminal at node 310 and a second terminalcoupled to the output node 304. The first inductor terminal is coupledto ground via the switch S5 (which may be referred to asde-magnetization switch) and to C_(F) via the first inductor switch S2at node 306. The second inductor terminal is coupled to the output node304.

A driver 320 is provided to generate a plurality of control signals Ct1,Ct2, Ct3, Ct4, Ct5 to operate the switches S1-S5 respectively. Thedriver 320 is adapted to operate the DC-DC converter 300 with a sequenceof states. The sequence of states may include a magnetization state tomagnetize the inductor and a de-magnetization state to de-magnetize theinductor. The driver may be configured to maintain the magnetizationstate and the de-magnetization state for a predetermined duration duringthe drive period. For instance, a duty cycle of the magnetization stateand a duty cycle of the de-magnetization state may be selected toachieve a target conversion ratio.

FIG. 4A illustrates the DC-DC converter of FIG. 3 operating in amagnetization state DP, in which the switches S2, and S4 are closedwhile the remaining switches S1, S3 and S5 are open. The input node 302is decoupled or disconnected from the output node 304. The ground iscoupled to the output node 304 via a path that includes the S4, C_(F),S2, and the inductor L.

FIG. 4B illustrates the DC-DC converter of FIG. 3 operating in ade-magnetization state DV, in which the switches S1, S3, and S5 areclosed while the remaining switches S2, and S4 are open. The input node302 is coupled to the output node 304 via an input path that includesC_(F) and S3 and bypasses the inductor L. The ground is coupled to theoutput node 304 via a de-magnetization path including S5 and theinductor L.

In operation the DC-DC power converter of FIG. 3 pulls no current fromthe input terminal during inductor magnetization (see FIG. 4A). Acurrent is pulled from the input terminal during the inductordemagnetization switching state (see FIG. 4B).

A ratio of average input to output currents during the duty cycle D_(V)of the de-magnetization state DV can be expressed as:

$\begin{matrix}\begin{matrix}{\frac{I_{IN}}{I_{OUT}} = \frac{D}{\left( {1 + D} \right)\left( {1 - D} \right)}} & {{{during}\mspace{14mu} D_{V}} - 1 - D} & {D \in \left\lbrack {0,1} \right\rbrack}\end{matrix} & (7)\end{matrix}$

In which D is the duty cycle of the magnetization state and D_(V) is theduty cycle of the de-magnetization state. For a duty cycle D<˜0.618(small output-to-input voltage conversion ratio) the level of the inputcurrent pulses I_(IN) is less than the level of the load currentI_(OUT).

The flying capacitor is automatically charged to V_(CF)=V_(IN)−V_(OUT)and the relationship between input and output voltage is obtained byapplying the volt-sec balance principle to the voltage of the inductor:

$\begin{matrix}\begin{matrix}{\frac{V_{OUT}}{V_{IN}} = \frac{D}{1 + D}} & {{D_{P} = D},{D_{V} = {1 - D}}} & {D \in \left\lbrack {0,1} \right\rbrack}\end{matrix} & (8)\end{matrix}$

In which D_(P) is the duty cycle of the magnetization state DP.

According to equation (8), the theoretical maximum converter voltageconversion ratio is V_(OUT)/V_(IN)=1/2 for D=1. However, for D=1,D_(V)=0 and there is no time available to re-distribute the charge fromthe flying capacitor C_(F) into the output capacitor C_(OUT) as thiswould require an infinite current causing a corresponding infinite I²Rconduction loss.

Current re-distribution may be achieved by selecting a duty cycle lessthan 1, for instance D≤3/4. For D=3/4 a practical maximum voltageconversion ratio of V_(OUT)/V_(IN)=3/7 is achieved.

For applications requiring a voltage conversion ratio greater thanV_(OUT)/V_(IN)=3/7 the inductor magnetization state DP of FIG. 4A may bereplaced or used in combination with a modified magnetization state DP2.

FIG. 4C illustrates the DC-DC converter of FIG. 3 operating in a secondmagnetization state DP2, in which the switches S1 and S2 are closedwhile the remaining switches S3, S4 and S5 are open. The input node 302is coupled to the output node 304 via a magnetization path that includesS1, S2 and the inductor L. The ground is not coupled to the output node304.

When introducing the second magnetization state DP2 into the drivingsequence at D>0.5, the relationship between input and output voltage maybe expressed as:

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{D}{2 - D}\mspace{31mu} D_{P2}} = {{2D} - 1}}},\mspace{14mu}{D_{P} = {D_{V} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {{0.5},1} \right\rbrack}}}} & (9)\end{matrix}$

In which D_(P2) is the duty cycle of the second magnetization state DP2.

By increasing the duty cycle D_(P2), the converter operationapproximates that of a traditional buck with an extended maximum dutycycle of D=1, a maximum voltage conversion ratio of V_(OUT)/V_(IN)=1 andthe amplitude of input current pulses approximating the level of theoutput current.

The efficiency of the DC-DC converter 300 may be improved for a lowoutput current by preventing a reverse output current. This can beachieved using a modified demagnetization state DV′.

FIG. 4D illustrates the DC-DC converter of FIG. 3 operating in a secondde-magnetization state DV′, in which the switches S1 and S3 are closedwhile the remaining switches S2, S4 and S5 are open. The input node 302is coupled to the output node 304 via an input path that includes S1,C_(F) and S3. The ground is not coupled to the output node 304.

The DC-DC converter 300 may be provided with a current sensor 330 (seeFIG. 3) for sensing an inductor current IL through the inductor. Thedriver 320 can be configured to operate the converter in a so-calledDiscontinuous Conduction Mode (DCM) in which the converter stopsprovision of current until the output voltage has dropped below athreshold value. In this example the driver 320 is configured to openthe de-magnetization switch S5 during the de-magnetization state DV uponsensing that the inductor current IL has reached a zero value. This maybe achieved using a zero-cross comparator circuit. Therefore, the driver320 drives the converter circuit with the second de-magnetization stateDV′ after the inductor current has fallen below zero. During themodified demagnetization state DV′ the current through the inductor isdiscontinued, however the current through the flying capacitor is stillprovided to the output port.

FIG. 5 illustrates a drive sequence for operating the DC-DC converter ofFIG. 3 with a conversion ratio

$\frac{V_{out}}{V_{in}} = {\frac{1}{12}.}$In this example, the driver 320 drives the DC-DC converter 300 with themagnetization state PD (waveform 510), between the times t0 and t1 for aduration Δ1, then with the de-magnetization state DV (waveform 520)between the time t1 and t2 for a duration Δ2. This sequence is thenrepeated over time to deliver the required output power. It will beappreciated that a delay also referred to as dead-time may be introducedat times t1 and t2.

For a voltage conversion ratio V_(OUT)/V_(IN)=1/12 the magnetizationduty cycle is D_(P)=1/11. As a result, the input current is flowingnearly continuously (more than 90% duty cycle) and its level is lessthan 10% of the load current as derived from equation (7). Therefore,the amplitude of input current pulses is more than 10 times lower thanthat of a conventional buck or of a 3-level buck converter. At smallvoltage conversion ratios, the input current approximates continuousconduction at input current levels that scale with the voltageconversion ratio.

For small step-down voltage conversion ratios (large step-up voltageconversion ratios) there is a relatively long duty cycle during whichthe current flows from the input terminal to the output terminal. Thisextended duty cycle of input current reduces the input current level,hence reducing the amplitude of pulsed current and associated voltageripples.

FIG. 6 is a diagram of another DC-DC converter 600 for implementing themethod of FIG. 2. The DC-DC converter 600 includes an inductor L and twoflying capacitors C_(F1) and C_(F2) coupled between a first port (inputnode 602) and a second port (output node 604) by a network of switchesformed of six switches S1-S6. An input capacitor Cin is provided betweenthe input node 602 and ground and an output capacitor Cout is providedbetween the output node 604 and ground.

The first flying capacitor C_(F1) has a first terminal coupled to theinput node 602 via the switch S5 (which may be referred to as inputswitch) and a second terminal coupled to ground via the ground switchS3. The second flying capacitor C_(F2) has a first terminal at node 606and a second terminal at node 608. The first terminal is coupled to theinput node 602 via a capacitor switch S1 and the input switch S5. Thesecond terminal is coupled to ground via the switch S4. The inductor Lhas a first terminal coupled to the second flying capacitor C_(F2) atnode 608 and a second terminal coupled to the output node 604. The firstterminal of C_(F2) is coupled to the output node 604 via switch S2. Theinductor second terminal is coupled to the second terminal of C_(F1) viaswitch S6. Therefore, the inductor L and the first and second flyingcapacitors C_(F1), C_(F2) are all connected to the output node 604,hence providing the option to split the output current across multipleparallel current paths.

A driver (not shown) is provided to generate six control signals Ct1-Ct6to operate the switches S1-S6 respectively. The driver is adapted tooperate the DC-DC converter 600 with a sequence of states. The sequenceof states may include a magnetization state and a de-magnetizationstate. The driver may be configured to maintain the magnetization stateand the de-magnetization state for a predetermined duration during thedrive period. For instance, a duty cycle of the magnetization state anda duty cycle of the de-magnetization state may be selected to achieve atarget conversion ratio.

FIG. 7A illustrates the DC-DC converter of FIG. 6 operating in amagnetization state DP, in which the switches S1 and S3 are closed whilethe remaining switches S2, S4, S5 and S6 are open. The input node 602 isdecoupled or disconnected from the output node 604. The ground iscoupled to the output node 604 via a path that includes S3, C_(F1), S1,C_(F2) and the inductor L.

FIG. 7B illustrates the DC-DC converter of FIG. 6 operating in ade-magnetization state DV, in which the switches S2, S4, S5 and S6 areclosed while the remaining switches S1 and S3 are open. The input node602 is coupled to the output node 604 via an input path that includesS5, C_(F1), S6, which bypasses the inductor L. The ground is coupled tothe output node 604 via two paths: a ground path and a de-magnetizationpath. The ground path includes S4, C_(F2), S2 while bypassing L. Thede-magnetization path includes S4 and the inductor L.

In operation the converter 600 automatically charges the flyingcapacitors to V_(CF2)=V_(OUT) and V_(CF1)=V_(IN)−V_(OUT) during thede-magnetization state DV and then connects C_(F1) and C_(F2) in seriesduring the magnetization state DP. The second flying capacitor increasesthe share of current bypassing the inductor. The topology of converter600 reduces further the amplitude of input current pulses, inductorcurrent and loss due to the inductor DC resistance.

A ratio of average input to output currents during the duty cycle D_(V)of the de-magnetization state DV can be expressed as:

$\begin{matrix}{\frac{I_{IN}}{I_{OUT}} = {{\frac{D}{\left( {1 + {2D}} \right)\left( {1 - D} \right)}\mspace{14mu}{during}\mspace{14mu} D_{V}} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {0,1} \right\rbrack}}} & (10)\end{matrix}$

The relationship between input and output voltage is expressed as:

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{D}{1 + {2D}}\mspace{31mu} D_{P}} = D}},\mspace{14mu}{D_{V} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {0,1} \right\rbrack}}} & (11)\end{matrix}$

For a voltage conversion ratio V_(OUT)/V_(IN)=1/12 the magnetizationduty cycle is D_(P)=1/10. The amplitude of input current pulses I_(IN)derived from equation (10) is just 5/54 of the load current I_(OUT).

The theoretical maximum voltage conversion ratio derived from equation(11) is V_(OUT)/V_(IN)=1/3 for D=1. However, for D=1, D_(V)=0 and thereis no time during the drive period to re-distribute the charge fromflying capacitors C_(F1) and C_(F2) into the output capacitor C_(OUT) asthis would require an infinite current causing a corresponding infiniteI²R conduction loss. Current distribution may be achieved by restrictingthe duty cycle to a value less than 1, for instance D≤3/4. For D=3/4 apractical maximum voltage conversion ratio of V_(OUT)/V_(IN)=3/10 isachieved.

Higher output voltages may be achieved by inserting a modifiedmagnetization state DP2 to the driving sequence.

FIG. 7C illustrates the DC-DC converter of FIG. 6 operating in a secondmagnetization state DP2, in which the switches S1 and S5 are closedwhile the remaining switches S2, S3, S4 and S6 are open. The ground nodeis decoupled from the output node 604. The input node 602 is coupled tothe output node 604 via a path that includes S5, S1, C_(F2) and theinductor L.

The second magnetization state DP2 introduces an inductor magnetizationcurrent from the input port through the second flying capacitor.Restricting the duty cycle to e.g. D≤3/4, increases the practicalmaximum voltage conversion ratio to V_(OUT)/V_(IN)=3/8 for D=3/4.

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{D}{2}\mspace{31mu} D_{P2}} = {{2D} - 1}}},{D_{P} = {D_{V} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {{0.5},{{0.7}5}} \right\rbrack}}}} & (12)\end{matrix}$

The topology of the converter of FIG. 6 improves conversion efficiencyby minimizing conduction losses in the converter and the externalcomponents (inductor DCR, capacitor ESR), but also by reducing inductorcore loss. In addition, the voltage drop from the flying capacitors alsoenables the use of switches, for instance power FETs, with a reducedvoltage rating. For example, the inductor de-magnetization switch S4 maybe implemented with a reduced voltage rating. This improves the figureof merit of the switches (smaller internal transistor resistance Ron andsmaller gate capacitance).

FIG. 8 is a diagram of another DC-DC converter 800 for implementing themethod of FIG. 2. The converter 800 can be implemented with powerswitches having a relatively low voltage rating for instance a voltagerating close to half the input voltage. The DC-DC converter 800 includesan inductor L and two flying capacitors C_(F1) and C_(F2) coupledbetween a first port (input node 802) and a second port (output node804) by a network of switches formed of eight switches S1-S8. An inputcapacitor Cin is provided between the input node 802 and ground and anoutput capacitor Cout is provided between the output node 804 andground.

The first flying capacitor C_(F1) has a first terminal at node 806coupled to C_(F2) via the capacitor switch S3 and a second terminal atnode 808 coupled to ground via the ground switch S5. The inductor L hasa first terminal at node 810 and a second terminal coupled to the outputnode 804. The first inductor terminal is coupled to ground via theswitch S8 (which may be referred to as de-magnetization switch) and toC_(F1) via the first inductor switch S6 at node 806. The second inductorterminal is coupled to the output node 804 and to C_(F1) via the secondcapacitor switch S7 at node 808. The second flying capacitor C_(F2) hasa first terminal at node 812 coupled to the input terminal via the inputswitch S1 and a second terminal at node 814 coupled to ground via theground switch S4. The first terminal of C_(F2) is coupled to the firstterminal of the inductor via the capacitor switch S2. A driver (notshown) is provided to generate eight control signals Ct1-Ct8 to operatethe switches S1-S8 respectively. The driver is adapted to operate theDC-DC converter 800 with a sequence of states. The sequence of statesmay include a magnetization state and a de-magnetization state. Thedriver may be configured to maintain the magnetization state and thede-magnetization state for a predetermined duration during the driveperiod. For instance, a duty cycle of the magnetization state and a dutycycle of the de-magnetization state may be selected to achieve a targetconversion ratio.

FIG. 9A illustrates the DC-DC converter of FIG. 8 operating in amagnetization state DP, in which the switches S2, S4, S5 and S6 areclosed while the remaining switches S1, S3, S7 and S8 are open. Theinput node 802 is decoupled or disconnected from the output node 804.The ground is coupled to the output node 804 via a first magnetizationpath that includes S5, C_(F1), S6 and the inductor L; and a secondmagnetization path that includes S4, C_(F2), S2 and the inductor L.

FIG. 9B illustrates the DC-DC converter of FIG. 8 operating in ade-magnetization state DV, in which the switches S1, S3, S7 and S8 areclosed while the remaining switches S2, S4, S5 and S6 are open. Theinput node 802 is coupled to the output node 804 via an input path thatincludes S1, C_(F2), S3, C_(F1) and that bypasses the inductor L. Theground is coupled to the output node 804 via a de-magnetization paththat includes S8 and the inductor L.

In operation the flying capacitors C_(F1) and C_(F2) are alternativelyconnected in series (during the de-magnetization state DV) and inparallel (during the magnetization state DP). The flying capacitorsC_(F1) and C_(F2) are automatically charged toV_(CF1)=V_(CF2)=(V_(IN)−V_(OUT))/2.

A ratio of average input to output currents during the duty cycle D_(V)of the de-magnetization state DV can be expressed as:

$\begin{matrix}{\frac{I_{IN}}{I_{OUT}} = {{\frac{D}{\left( {2 + D} \right)\left( {1 - D} \right)}\mspace{14mu}{during}\mspace{14mu} D_{V}} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {0,1} \right\rbrack}}} & (13)\end{matrix}$

The relationship between input and output voltage is expressed as:

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{D}{2 + D}\mspace{31mu} D_{P}} = D}},\mspace{14mu}{D_{V} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {0,1} \right\rbrack}}} & (14)\end{matrix}$

By combining equations 13 and 14

$\frac{I_{IN}}{I_{OUT}} = {\frac{V_{OUT}}{V_{IN}}{\frac{1}{\left( {1 - D} \right)}.}}$As a result, the level of input current scales with the product of thevoltage conversion ratio with 1/(1-D).

For a voltage conversion ratio V_(OUT)/V_(IN)=1/12 the magnetizationduty cycle is D_(P)=2/11. The amplitude of input current pulses I_(IN)derived from equation (13) is just ˜10% (11/108) of the load currentI_(OUT).

The theoretical maximum voltage conversion ratio derived from equation(14) is V_(OUT)/V_(IN)=1/3 for D=1. However, for D=1, there is no timeduring the drive period D_(V)=0 to re-distribute the charge from flyingcapacitors C_(F1) and C_(F2) into the output capacitor C_(OUT) as thiswould require an infinite current causing a corresponding infinite I²Rconduction loss. Current distribution may be achieved by restricting theduty cycle to a value less than 1, for instance D≤3/4. For D=3/4 apractical maximum voltage conversion ratio of V_(OUT)/V_(IN)=3/11 isachieved.

FIG. 10 is a diagram of another DC-DC converter 1000 for implementingthe method of FIG. 2. The converter 1000 is designed for operating atvery small voltage conversion ratios, for instance for Vout/Vin<1/7. Forexample a typical conversion ratio may be Vout/Vin=1/12. The converter1000 is similar to the converter 600, in which the switch S5 has beenreplaced by a pre-converter stage. The pre-converter stage, alsoreferred to as first port stage or input stage, may be implemented as aserial-parallel topology, a Dickson topology or any other capacitivevoltage divider topology.

The DC-DC converter 1000 includes an inductor L and three flyingcapacitors C_(F1), C_(F2) and C_(F3) coupled between a first port (inputnode 1002) and a second port (output node 1004) by a network of switchesformed of nine switches S1-S9. An input capacitor Cin is providedbetween the input node 1002 and ground and an output capacitor Cout isprovided between the output node 1004 and ground.

An input stage is provided between the input node 1002 and anintermediate node 1014, and an output stage is provided between theintermediate node 1014 and the output node 1004. The input stage isformed of C_(F3) and switches S1, S2, S3, S4. The third flying capacitorC_(F3) has a first terminal at node 1010 coupled to the input node viathe input switch S1 and a second terminal at node 1012 coupled to groundvia the ground switch S4. The first terminal of C_(F3) is coupled to theoutput stage via the capacitor switch S2 provided between the nodes 1010and 1014. Similarly the second terminal of C_(F3) is coupled to theoutput stage via the capacitor switch S3 provided between the nodes 1012and 1014.

The output stage is formed of C_(F1), C_(F2) and switches S5, S6, S7,S8, and S9. The first flying capacitor C_(F1) has a first terminalcoupled to the second flying capacitor C_(F2) via the switch S5 and asecond terminal coupled to ground via the ground switch S7, and to theoutput node 1004 via switch S9. The second flying capacitor C_(F2) has afirst terminal at node 1006 coupled to S5 and a second terminal at node1008 coupled to ground via S8. The inductor L has a first terminalcoupled to the second flying capacitor C_(F2) at node 1008 and a secondterminal coupled to the output node 1004. The first terminal of C_(F2)is coupled to the output node 1004 via switch S6. A driver (not shown)is provided to generate nine control signals Ct1-Ct9 to operate theswitches S1-S9 respectively. The driver is adapted to operate the DC-DCconverter 1000 with a sequence of states. The sequence of states mayinclude a magnetization state and a de-magnetization state. The drivermay be configured to maintain the magnetization state and thede-magnetization state for a predetermined duration during the driveperiod. For instance, a duty cycle of the magnetization state and a dutycycle of the de-magnetization state may be selected to achieve a targetconversion ratio.

FIG. 11A illustrates the DC-DC converter of FIG. 10 operating in amagnetization state D1, in which the switches S5 and S7 are closed whilethe switches S2, S4, S6, S8 and S9 are open, and at least one of S1 andS3 are also open. The input node is decoupled or disconnected from theoutput node. The ground is coupled to the output node via a path thatincludes S7, C_(F1), S5, C_(F2) and the inductor L.

FIG. 11B illustrates the DC-DC converter of FIG. 10 operating in ade-magnetization state DV1, in which the switches S1, S3, S6, S8 and S9are closed while the remaining switches S2, S4, S5 and S7 are open. Theinput node is coupled to the output node via an input path that includesS1, C_(F3), S3, C_(F1), S9 which bypasses the inductor L. The ground iscoupled to the output node via two paths: a ground path and ade-magnetization path. The ground path includes S8, C_(F2), S6 whilebypassing L. The de-magnetization path includes S8 and the inductor L.

FIG. 11C illustrates the DC-DC converter of FIG. 10 operating in asecond magnetization state D2, in which the switches S5 and S7 areclosed while the switches S1, S3, S6, S8 and S9 are open, and at leastone of S2 and S4 are also open. The input node is decoupled ordisconnected from the output node. The ground is coupled to the outputnode via a path that includes S7, C_(F1), S5, C_(F2) and the inductor L.

FIG. 11D illustrates the DC-DC converter of FIG. 10 operating in ade-magnetization state DV2, in which the switches S2, S4, S6, S8 and S9are closed while the remaining switches S1, S3, S5 and S7 are open. Theinput node is de-coupled from the output node. The ground is coupled tothe output node via three paths: a first ground path, a second groundpath and a de-magnetization path. The first ground path includes S4,C_(F3), S2, C_(F1), S9 and bypasses the inductor L. The second groundpath includes S8, C_(F2), S6 while bypassing L. The de-magnetizationpath includes S8 and the inductor L.

FIG. 12 illustrates a drive sequence for operating the DC-DC converter1000. The drive sequence has a drive period T=T1+T2, in which T1 is thedrive period of a cycle of the states D1 and DV1 and T2, in which T2 isthe drive period of a cycle of the states D2 and DV2. In this example,the driver drives the DC-DC converter 1000 with the magnetization stateD1 (waveform 1210) between the times t0 and t1 for a duration Δ1, thenwith the de-magnetization state DV1 (waveform 1220) between the time t1and t2 for a duration Δ2, then with the magnetization state D2 (waveform1230) between the times t2 and t3 for a duration Δ3, then with thede-magnetization state DV2 (waveform 1240) between the time t3 and t4for a duration Δ4. This drive sequence D1/DV1/D2/DV2 is then repeatedover time to deliver the required output power.

In operation the flying capacitors are automatically charged towardsV_(CF3)=V_(IN)/2, V_(CF2)=V_(OUT) and V_(CF1)=V_(IN)/2−V_(OUT).

The ratio between input and load current level follows equation (10).The relation between input and output voltage may be expressed as:

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{D}{2 + {4D}}\mspace{31mu}{Dx}} = D}},\mspace{14mu}{{DVx} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {0,1} \right\rbrack}}} & (15)\end{matrix}$

in which D_(X) is the duty cycle of the magnetization state D1 or D2 andD_(VX) is the duty cycle of the de-magnetization state DV1 or DV2.

The theoretical maximum voltage conversion ratio derived from equation(15) is V_(OUT)/V_(IN)=1/4 for D=1. However, for D=1, D_(VX)=0 and thereis no time during the drive period to re-distribute the charge fromflying capacitors C_(F1) and C_(F2) into the output capacitor C_(OUT) asthis would require an infinite current causing a corresponding infiniteI²R conduction loss. The charge of capacitor C_(F3) is controlled by theratio DV1/DV2. Current distribution may be achieved by restricting theduty cycle to a value less than 1, for instance D≤3/4. For D=3/4 apractical maximum voltage conversion ratio of V_(OUT)/V_(IN)=3/20 isachieved. The drive sequence of FIG. 12 is illustrated for a conversionratio

$\frac{V_{out}}{V_{in}} = \frac{3}{20}$in which T1=T2=T/2, Δ1=Δ3=3/4 T1, and Δ2=Δ4=1/4 T1.

Higher output voltages may be achieved by using other states (in thecase of a Buck converter, additional magnetizations states) in additionto the switching states D1 and D2.

FIG. 13A illustrates the DC-DC converter of FIG. 10 operating in amagnetization state DP1, in which the switches S1, S3, S5 are closedwhile the remaining switches S2, S4, S6, S7, S8 and S9 are open. Theinput node is coupled to the output node via a magnetization path thatincludes S1, C_(F3), S3, S5, C_(F2) and L. The ground is de-coupled fromthe output node.

FIG. 13B illustrates the DC-DC converter of FIG. 10 operating in amagnetization state DP2, in which the switches S2, S4 and S5 are closedwhile the remaining switches S1, S3, S6, S7, S8 and S9 are open. Theinput node is decoupled or disconnected from the output node. The groundis coupled to the output node via a path that includes S4, C_(F3), S2,S5, C_(F2) and the inductor L.

These states introduce inductor magnetization current from the inputport (through flying capacitor C_(F2)). By restricting the duty cycle toD≤3/4, an increased practical maximum voltage conversion ratio can beachieved. For example, for D=3/4, the ratio V_(OUT)/V_(IN)=3/16.

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{D}{4}\mspace{31mu} D_{P2}} = {{2D} - 1}}},{D_{P} = {D_{V} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {{0.5},{{0.7}5}} \right\rbrack}}}} & (16)\end{matrix}$

FIG. 14 is a flow chart of another method for converting power with atarget conversion ratio according to the disclosure.

At step 1410 a power converter having a ground port, a first port, and asecond port is provided. The power converter can operate either as astep-down converter or as a step-up converter. When the power converteroperates as a step-down converter the first port receives an inputvoltage and the second port provides the output voltage. When the powerconverter operates as a step-up converter the second port receives aninput voltage and the first port provides the output voltage. The powerconverter includes a first flying capacitor coupled to a network ofswitches, a second flying capacitor coupled to the network of switches,an inductor coupled to the second port, and a driver. The network ofswitches comprises a first switch to couple the second flying capacitorto the first port; a ground switch to couple the inductor to ground, anda first capacitor switch coupled to the first flying capacitor.

At step 1420, the network of switches is driven with a sequence ofstates during a drive period. The sequence of states comprises a firststate and a second state. In the first state the ground port is coupledto the second port via a first path comprising the first flyingcapacitor and the inductor, and wherein the first port is coupled to thesecond port via a second path comprising the first switch, the secondflying capacitor and the inductor. In the second state the ground portis coupled to the second port via a third path comprising the groundswitch and the inductor, and one of the first port and the ground portis coupled to the second port via a fourth path comprising the firstflying capacitor while bypassing the inductor. As a result, in the firststate a reduced current is flowing between the first port and the secondport. For instance, when operating as a buck converter, a reducedcurrent is pulled from the first port to the second port. Similarly,when operating as a boost converter, a reduced current is pulled fromthe second port to the first port.

When the power converter operates as a step-down converter, the firststate is a magnetization state and the second state is ade-magnetization state. Conversely, when the power converter operates asa step-up converter, the first state is a de-magnetization state and thesecond state is a magnetization state.

FIG. 15 is a diagram of a DC-DC converter 1500 for implementing themethod of FIG. 14. The DC-DC converter 1500 includes an inductor L andtwo flying capacitors C_(F1) and C_(F2) coupled between a first port(input node 1502) and a second port (output node 1504) by a network ofswitches formed of seven switches S1-S7. An input capacitor Cin isprovided between the input node 1502 and ground and an output capacitorCout is provided between the output node 1504 and ground.

The first flying capacitor C_(F1) has a first terminal at node 1506coupled to the input node 1502 via the capacitor switch S5 and a secondterminal at node 1508 coupled to ground via the ground switch S3. Thesecond flying capacitor C_(F2) has a first terminal at node 1510 coupledto the input node via the switch S1 (also referred to as input switch)and a second terminal at node 1512 coupled to ground via the groundswitch S4. The inductor L has a first terminal at node 1512 and a secondterminal coupled to the output node 1504. The first inductor terminal iscoupled to C_(F1) via the inductor switch S6 at node 1506, and to C_(F2)at node 1512. The first inductor terminal is also coupled to ground viathe switch S4. The second terminal of C_(F1) is coupled to the outputnode 1504 via the switch S7. The first terminal of C_(F2) is coupled tothe output node 1504 via the switch S2. A driver (not shown) is providedto generate seven control signals Ct1-Ct7 to operate the switches S1-S7respectively. The driver is adapted to operate the DC-DC converter 1500with a sequence of states. The sequence of states may include amagnetization state and a de-magnetization state. The driver may beconfigured to maintain the magnetization state and the de-magnetizationstate for a predetermined duration during the drive period. Forinstance, a duty cycle of the magnetization state and a duty cycle ofthe de-magnetization state may be selected to achieve a targetconversion ratio.

FIG. 16A illustrates the DC-DC converter of FIG. 15 operating in amagnetization state DP, in which the switches S1, S3 and S6 are closedwhile the remaining switches S2, S4, S5 and S7 are open. The input node1502 is coupled to the output node 1504 via a first path ormagnetization path that includes S1, C_(F2) and the inductor L. Theground port is coupled to the output node 1504 via a second path orsecond magnetization path that includes S3, C_(F1), S6 and the inductorL.

FIG. 16B illustrates the DC-DC converter of FIG. 15 operating in ade-magnetization state DV, in which the switches S2, S4, S5 and S7 areclosed while the remaining switches S1, S3 and S6 are open. The inputnode 1502 is coupled to the output node 1504 via a path that includesS5, C_(F1), S7, which bypasses the inductor L. The ground is coupled tothe output node 1504 via a path also referred to as de-magnetizationpath that includes S4 and the inductor L; and by another path alsoreferred to as ground path that includes S4, C_(F2), and S2, which alsobypasses the inductor L.

In operation the flying capacitors are automatically charged toV_(CF2)=V_(OUT) and V_(CF1)=V_(IN)−V_(OUT). The relationship betweeninput and output voltage follows equation (8).

As illustrated in FIGS. 16A and 16B a current is provided from the inputterminal during both the magnetization state DP and the de-magnetizationstate DV. As a result, the converter 1500 implements continuous inputcurrent over the driving period and the input current is reducedcompared with the converter 600 of FIG. 6 especially for voltageconversion ratios close to the maximum ratio of V_(OUT)/V_(IN)=1/2. Forthis range of operation, the converter 1500 operates like a transformerfor DC voltages with an input current level close to 1/2 of the loadcurrent.

FIG. 17 is a diagram of another DC-DC converter 1700 for implementingthe method of FIG. 14. The DC-DC converter 1700 includes an inductor Land two flying capacitors C_(F1) and C_(F2) coupled between a first port(input node 1702) and a second port (output node 1704) by a network ofswitches formed of six switches S1-S6. An input capacitor Cin isprovided between the input node 1702 and ground and an output capacitorCout is provided between the output node 1704 and ground.

The first flying capacitor C_(F1) has a first terminal at node 1706coupled to C_(F2) via the capacitor switch S5, and a second terminal atnode 1708 coupled to ground via the ground switch S3, and to the outputnode 1704 via switch S6.

The second flying capacitor C_(F2) has a first terminal at node 1710coupled to the input node 1702 via the first switch or input switch S1,and a second terminal at node 1712 coupled to ground via the groundswitch S4. The inductor L has a first inductor terminal coupled toground via S4 and a second inductor terminal coupled to the output node1704. The first inductor terminal is coupled to C_(F2) at node 1712, andto C_(F1) via the inductor switch S2 at node 1706. A driver (not shown)is provided to generate six control signals Ct1-Ct6 to operate theswitches S1-S6 respectively. The driver is adapted to operate the DC-DCconverter 1700 with a sequence of states. The sequence of states mayinclude a magnetization state and a de-magnetization state. The drivermay be configured to maintain the magnetization state and thede-magnetization state for a predetermined duration during the driveperiod. For instance, a duty cycle of the magnetization state and a dutycycle of the de-magnetization state may be selected to achieve a targetconversion ratio.

FIG. 18A illustrates the DC-DC converter of FIG. 17 operating in amagnetization state DP, in which the switches S1, S2, and S3 are closedwhile the remaining switches S4, S5 and S6 are open. The input node 1702is coupled to the output node 1704 via a magnetization path thatincludes S1, C_(F2) and the inductor L. The ground is coupled to theoutput node 1704 via a path that includes the S3, C_(F1), S2 and theinductor L.

FIG. 18B illustrates the DC-DC converter of FIG. 17 operating in ade-magnetization state DV, in which the switches S4, S5, and S6 areclosed while the remaining switches S1, S2, and S3 are open. The inputnode 1702 is de-coupled from the output node 1704. The ground is coupledto the output node 1704 via a de-magnetization path including S4 and theinductor L, and via another path including S4, C_(F2), S5, C_(F1) andS6, which bypasses the inductor L.

During the magnetization state DP, the converter 1700 typically provideshalf of the inductor magnetization current from the input terminal (viaflying capacitor C_(F2)) and the other half from the ground terminal(via flying capacitor C_(F1)). During the de-magnetization state, theflying capacitors are connected in series to provide a supplement outputcurrent from the ground terminal. This operation prevents the occurrenceof current spikes from the input node, typically generated whenconnecting the input and output capacitors directly through a flyingcapacitor.

The flying capacitors are automatically charged toV_(CF2)=(V_(IN)+V_(OUT))/2 and V_(CF1)=(V_(IN)−V_(OUT))/2.

A ratio of average input to output currents during the duty cycle D_(P)of the magnetization state DP can be expressed as:

$\begin{matrix}{\frac{I_{IN}}{I_{OUT}} = {{\frac{1}{2 + D}\mspace{14mu}{during}\mspace{14mu} D_{P}} = {{D\mspace{31mu} D} \in \left\lbrack {0,1} \right\rbrack}}} & (17)\end{matrix}$

The relationship between input and output voltage is:

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{D}{2 + D}\mspace{31mu} D_{P}} = D}},\mspace{14mu}{D_{V} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {0,1} \right\rbrack}}} & (18)\end{matrix}$

For a voltage conversion ratio V_(OUT)/V_(IN)=1/4 the duty cycle isD=2/3. The amplitude of input current pulses I_(IN) derived fromequation (17) is just 3/8 of the load current I_(OUT).

The theoretical maximum voltage conversion ratio derived from equation(18) is V_(OUT)/V_(IN)=1/3 for D=1. However, for D=1, D_(V)=0 and thereis no time available during the drive period to re-distribute the chargefrom flying capacitors C_(F1) and C_(F2) into the output capacitorC_(OUT) as this would require an infinite current causing acorresponding infinite I²R conduction loss. Current distribution may beachieved by restricting the duty cycle to a value less than 1, forinstance D≤3/4. For D=3/4 a practical maximum voltage conversion ratioof V_(OUT)/V_(IN)=3/11 is achieved.

Therefore for output-to-input voltage conversion ratios larger thanV_(OUT)/V_(IN)=1/4, the converter 1700 reduces the amplitude of inputcurrent pulses and reduces also the voltage rating of thedemagnetization switch to approximately half the maximum input voltage.

The converter 1700 may be modified by replacing the switch S6 by a fixedconnection between the second terminal of C_(F1) and the output node andby removing the ground connection of C_(F1) via S3. In this case thevoltages across the flying capacitors would be expressed asV_(CF2)=V_(IN)/2 and V_(CF1)=V_(IN)/2−V_(OUT). However in this scenariothe output current during inductor magnetization is reduced to ˜50% ofthe inductor current, resulting in a slower transient load response andan increase in output current/voltage ripple especially at high dutycycle. The converter 800 may also be modified in a similar fashion.

The DC-DC converters described in relation to FIGS. 3 to 18 areconfigured to reduce the amplitude of input current pulses compared withconventional converters. This reduces both the power losses and thenoise level on the power supply and corresponding EMI issues. Byimplementing a capacitive path bypassing the inductor, the losses due tothe inductor DCR can also be reduced hence improving converterefficiency, voltage regulation and improving response to transient loadcurrent. Furthermore, when ramping-up the inductor current via anextended magnetization state, an additional charge is stored into theflying capacitors. This charge is consequently provided to the converteroutput port during the consecutive demagnetization state. This furtherreduces the output voltage drop during a sudden rise in load current. Inaddition, the voltage across the flying capacitor(s) does not requireany regulation, hence reducing complexity in the control circuitry andthe risk of interference with the regulation loops of converter outputvoltage and current.

The DC-DC converters described in relation to FIGS. 3 to 18 have beendescribed as step-down converters also referred to as Buck converters.It will be appreciated that these converters may be operated in reverse(that is using the input as the output and the output as the input) asBoost converters to achieve step-up conversion. In this case, themagnetization (de-magnetization) phase in the buck operation becomes ade-magnetization (magnetization) phase in the boost operation.

The transfer function of a traditional boost converter contains aso-called right-half-plane zero, as described in publication titled“Right-Half-Plane Zero Elimination for Boost Converter Using MagneticCoupling With Forward Energy Transfer”, IEEE, 2019 by Poorali. The zeroresults from the fact that a converter provides the output currentduring inductor demagnetization. This limits the bandwidth of aclosed-loop control system in continuous conduction mode (CCM). As aresult traditional boost converters are implementing increased outputvoltage ripple for applications having fast dynamics.

FIG. 19 shows the diagram of FIG. 3 represented with inverted input andoutput ports.

FIGS. 20A and 20B illustrate the magnetization state DP and thedemagnetization state DV, respectively.

FIG. 20A shows the DC-DC converter of FIG. 19 operating in amagnetization state DP, in which the switches S1, S3, and S5 are closedwhile the remaining switches S2, and S4 are open. The input node iscoupled to the output node via an input path that includes C_(F) and S3and bypasses the inductor L. The input node is coupled to ground via amagnetization path including S5 and the inductor L.

FIG. 20B illustrates the DC-DC converter of FIG. 19 operating in ade-magnetization state DV, in which the switches S2, and S4 are closedwhile the remaining switches S1, S3 and S5 are open. The input node isdecoupled or disconnected from the output node. The input node iscoupled to the ground via a de-magnetization path that includes the S4,C_(F), S2, and the inductor L.

In operation the DC-DC power converter of FIG. 19 pulls no current fromthe input terminal during inductor de-magnetization (see FIG. 20B). Acurrent is pulled from the input terminal during the inductormagnetization switching state (see FIG. 20A). As illustrated above withrespect to FIGS. 20A and 20B, the proposed topologies of the disclosuretransfer the provision of converter output current into the switchingstate that magnetizes the inductor, effectively shifting theright-half-plane zero from the transfer function of the boost controlloop to higher frequency.

Compared with transformer-less converters of the prior art, theconverter topologies of the disclosure enable large voltage ratio boostconversion with improved power supply rejection and fast dynamicresponse.

The relationship between input and output voltage is obtained byapplying the volt-sec balance principle to the voltage of the inductor:

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{2 - D}{1 - D}\mspace{31mu} D_{P}} = D}},\mspace{14mu}{D_{V} = {{1 - {D\mspace{31mu} D}} \in \left\lbrack {0,1} \right\rbrack}}} & (19)\end{matrix}$

According to equation (19), the theoretical minimum converter voltageconversion ratio is V_(OUT)/V_(IN)=2 for D=0. However, for D=0 there isno time available to re-distribute the charge from flying capacitorC_(F) into the output capacitor COUT (this would require an infinitecurrent causing a corresponding infinite I²R conduction loss). A morebalanced current distribution may be achieved by restricting the dutycycle to e.g. D≥1/4, resulting in a more realistic minimum voltageconversion ratio of V_(OUT)/V_(IN)=7/3 for D=1/4. For lower voltageconversion ratios, the switching state DV may be replaced partially orentirely with a modified demagnetization state DV2.

FIG. 21 illustrates the DC-DC converter of FIG. 19 operating in a secondde-magnetization state DV2, in which the switches S1 and S2 are closedwhile the remaining switches S3, S4 and S5 are open. The input node iscoupled to the output node via a de-magnetization path that includes S1,S2 and the inductor L. The ground is not coupled to the output node.

By introducing an increasing share of DV2 for duty cycles below D<0.5the relationship between input and output voltages becomes:

$\begin{matrix}{{\frac{V_{OUT}}{V_{IN}} = {{\frac{1 + D}{1 - D}\mspace{31mu} D_{P}} = {D_{V} = D}}},{{DV2} = {{1 - {2D\mspace{31mu} D}} \in \left\lbrack {0,{0.5}} \right\rbrack}}} & (20)\end{matrix}$

An increasing share of switching state DV2 during inductordemagnetization makes the converter operation similar to that of atraditional boost converter with a minimum duty cycle of D=0 and aminimum voltage conversion ratio of V_(OUT)/V_(IN)>1. This has also thedrawback of re-introducing larger impact from the right-half-plane zero.

Disabling negative inductor current at low output current to increaseconverter efficiency may be applied to step-up derivatives of theproposed converter topologies by opening the demagnetizing current pathwithin the demagnetizing state D_(VX) as soon as the inductor current isreaching zero.

Reducing the voltage rating of boost converter power switches toV_(OUT)/2 may be achieved for the topologies of FIGS. 8, 10 and 17 withinverted roles of input and output ports.

A skilled person will appreciate that variations of the disclosedarrangements are possible without departing from the disclosure.Accordingly, the above description of the specific embodiment is made byway of example only and not for the purposes of limitation. It will beclear to the skilled person that minor modifications may be made withoutsignificant changes to the operation described.

What is claimed is:
 1. A power converter for providing an output voltagewith a target conversion ratio, the power converter having a groundport, a first port, and a second port, wherein when the power converteroperates as a step-down converter the first port receives an inputvoltage and the second port provides the output voltage and when thepower converter operates as a step-up converter the second port receivesan input voltage and the first port provides the output voltage, thepower converter comprising a first flying capacitor coupled to a networkof switches, a second flying capacitor coupled to the network ofswitches, an inductor coupled to the second port, and a driver; thenetwork of switches comprising a first switch to couple the secondflying capacitor to the first port; a ground switch to couple theinductor to ground; a first capacitor switch coupled to the first flyingcapacitor; the driver being adapted to drive the network of switcheswith a sequence of states during a drive period, the sequence of statescomprising a first state and a second state, wherein in the first statethe ground port is coupled to the second port via a first pathcomprising the first flying capacitor and the inductor, and wherein thefirst port is coupled to the second port via a second path comprisingthe first switch, the second flying capacitor and the inductor, whereinin the second state the ground port is coupled to the second port via athird path comprising the ground switch and the inductor, and whereinone ofthe first port or the ground port is coupled to the second portvia a fourth path comprising the first flying capacitor while bypassingthe inductor.
 2. The power converter as claimed in claim 1, wherein inthe second state the first port is decoupled from the second port andwherein the fourth path comprises the second flying capacitor.
 3. Thepower converter as claimed in claim 1, wherein in the second state theground port is coupled to the second port via a fifth path comprisingthe second flying capacitor, while bypassing the inductor.
 4. The powerconverter as claimed in claim 1, wherein the inductor has a firstterminal coupled to the first flying capacitor via a first inductorswitch, and a second terminal connected to the second port, and whereinthe first flying capacitor is coupled to the second port via a secondcapacitor switch.
 5. The power converter as claimed in claim 4, whereinthe second flying capacitor is coupled to the second port via a thirdcapacitor switch.
 6. The power converter as claimed in claim 1, furthercomprising a current sensor for sensing an inductor current through theinductor, wherein the driver is adapted to open the ground switch duringthe second state upon sensing that the inductor current has reached athreshold value.
 7. The power converter as claimed in claim 1, whereinthe power converter is a step-down converter, the first state being amagnetization state and the second state being a de-magnetization state.8. The power converter as claimed in claim 1, wherein the powerconverter is a step-up converter, the first state being ade-magnetization state and the second state being a magnetization state.9. A method of converting power with a target conversion ratio, themethod comprising providing a power converter having a ground port, afirst port, and a second port, wherein when the power converter operatesas a step-down converter the first port receives an input voltage andthe second port provides the output voltage and when the power converteroperates as a step-up converter the second port receives an inputvoltage and the first port provides the output voltage, the powerconverter further comprising a first flying capacitor coupled to anetwork of switches, a second flying capacitor coupled to the network ofswitches, an inductor coupled to the second port, and a driver; whereinthe network of switches comprises a first switch to couple the secondflying capacitor to the first port; a ground switch to couple theinductor to ground; a first capacitor switch coupled to the first flyingcapacitor; driving the network of switches with a sequence of statesduring a drive period, the sequence of states comprising a first stateand a second state, wherein in the first state the ground port iscoupled to the second port via a first path comprising the first flyingcapacitor and the inductor, and wherein the first port is coupled to thesecond port via a second path comprising the first switch, the secondflying capacitor and the inductor, wherein in the second state theground port is coupled to the second port via a third path comprisingthe ground switch and the inductor, and wherein one ofthe first port orthe ground port is coupled to the second port via a fourth pathcomprising the first flying capacitor while bypassing the inductor. 10.The method as claimed in claim 9, wherein in the second state the firstport is decoupled from the second port and wherein the fourth pathcomprises the second flying capacitor.
 11. The method as claimed inclaim 9, wherein in the second state the ground port is coupled to thesecond port via a fifth path comprising the second flying capacitor,while bypassing the inductor.